A common signal format for mobile wireless communications is orthogonal frequency-domain multiplexing (OFDM) (see, for example, en.wikipedia.org/Orthogonal_frequency-division_multiplexing), and closely related formats such as orthogonal frequency-domain multiple access (OFDMA). For a signal conveyed on an OFDM channel, this is characterized in the frequency domain by a bundle of narrow adjacent subchannels, and in the time domain by a relatively slow series of OFDM symbols each with a time T, each separated by a guard interval ΔT (see FIG. 1B). Within the guard interval before each symbol is a cyclic prefix (CP), comprised of the same signal in the symbol period, cyclically shifted in time. This CP is designed to reduce the sensitivity of the received signal to precise time synchronization in the presence of multipath, i.e., radio-frequency signals reflecting from large objects in the terrain such as tall buildings, hills, etc. If a given symbol is received with a slight time delay (less than ΔT), it will still be received without error. In addition to the data symbols associated with the OFDM “payload”, there is also typically a “preamble” signal that establishes timing and other standards. The preamble may have its own CP, not shown in FIG. 1B.
In addition to the preamble, a set of pilot symbols (also called training symbols) are typically interleaved (in time and frequency) among the data symbols in the payload. These pilot symbols are used together with the preamble for further refinement of timing, channel estimation, and signal equalization at the receiver. The particular placement of pilot symbols in time and frequency within the payload may differ among various OFDM standard protocols. A typical example of the placement of pilot symbols in the time-frequency resource grid is shown in FIG. 2 for a protocol known as “Long-Term Evolution” (LTE). (See, for example, www.mathworks.com/help/lte/ug/channel-estimation.html for further information.) Here pilot symbols are located at four different frequencies, with a pattern that repeats every eight symbol periods. This enables the receiver to obtain information on time-varying channel estimation across the entire resource grid, using interpolation of the various pilot symbols.
In OFDM, the sub-carrier frequencies are chosen so that the sub-carriers are orthogonal to each other, meaning that cross-talk between the sub-channels is eliminated and inter-sub-carrier guard bands are not required. This greatly simplifies the design of both the transmitter and the receiver; unlike conventional FDM, a separate filter for each sub-channel is not required. The orthogonality requires that the sub-carrier spacing is Δf=k/(TU) Hertz, where TU seconds is the useful symbol duration (the receiver side window size), and k is a positive integer, typically equal to 1. Therefore, with N sub-carriers, the total passband bandwidth will be B≈N·Δf (Hz).
The orthogonality also allows high spectral efficiency, with a total symbol rate near the Nyquist rate. Almost the whole available frequency band can be utilized. OFDM generally has a nearly “white” spectrum, giving it benign electromagnetic interference properties with respect to other co-channel users.
When two OFDM signals are combined, the result is in general a non-orthogonal signal. While a receiver limited to the band of a single OFDM signal would be generally unaffected by the out-of-channel signals, when such signals pass through a common power amplifier, there is an interaction, due to the inherent non-linearities of the analog system components.
OFDM requires very accurate frequency synchronization between the receiver and the transmitter; with frequency deviation the sub-carriers will no longer be orthogonal, causing inter-carrier interference (ICI), i.e. cross-talk between the sub-carriers. Frequency offsets are typically caused by mismatched transmitter and receiver oscillators, or by Doppler shift due to movement. While Doppler shift alone may be compensated for by the receiver, the situation is worsened when combined with multipath, as reflections will appear at various frequency offsets, which is much harder to correct.
The orthogonality allows for efficient modulator and demodulator implementation using the fast Fourier transform (FFT) algorithm on the receiver side, and inverse FFT (IFFT) on the sender side. While the FFT algorithm is relatively efficient, it has modest computational complexity which may be a limiting factor.
One key principle of OFDM is that since low symbol rate modulation schemes (i.e. where the symbols are relatively long compared to the channel time characteristics) suffer less from intersymbol interference caused by multipath propagation, it is advantageous to transmit a number of low-rate streams in parallel instead of a single high-rate stream. Since the duration of each symbol is long, it is feasible to insert a guard interval between the OFDM symbols, thus eliminating the intersymbol interference. The guard interval also eliminates the need for a pulse-shaping filter, and it reduces the sensitivity to time synchronization problems.
The cyclic prefix, which is transmitted during the guard interval, consists of the end of the OFDM symbol copied into the guard interval, and the guard interval is transmitted followed by the OFDM symbol. The reason that the guard interval consists of a copy of the end of the OFDM symbol is so that the receiver will integrate over an integer number of sinusoid cycles for each of the multipaths when it performs OFDM demodulation with the FFT.
The effects of frequency-selective channel conditions, for example fading caused by multipath propagation, can be considered as constant (flat) over an OFDM sub-channel if the sub-channel is sufficiently narrow-banded, i.e. if the number of sub-channels is sufficiently large. This makes equalization far simpler at the receiver in OFDM in comparison to conventional single-carrier modulation. The equalizer only has to multiply each detected sub-carrier (each Fourier coefficient) by a constant complex number, or a rarely changed value. Therefore, receivers are generally tolerant of such modifications of the signal, without requiring that explicit information be transmitted.
OFDM is invariably used in conjunction with channel coding (forward error correction), and almost always uses frequency and/or time interleaving. Frequency (subcarrier) interleaving increases resistance to frequency-selective channel conditions such as fading. For example, when a part of the channel bandwidth is faded, frequency interleaving ensures that the bit errors that would result from those subcarriers in the faded part of the bandwidth are spread out in the bit-stream rather than being concentrated. Similarly, time interleaving ensures that bits that are originally close together in the bit-stream are transmitted far apart in time, thus mitigating against severe fading as would happen when travelling at high speed. Therefore, similarly to equalization per se, a receiver is typically tolerant to some degree of modifications of this type, without increasing the resulting error rate.
The OFDM signal is generated from the digital baseband data by an inverse (fast) Fourier transform (IFFT), which is computationally complex, and as will be discussed below, generates a resulting signal having a relatively high peak to average power ratio (PAPR) for a set including a full range of symbols. This high PAPR, in turn generally leads to increased acquisition costs and operating costs for the power amplifier (PA), and typically a larger non-linear distortion as compared to systems designed for signals having a lower PAPR. This non-linearity leads, among other things, to clipping distortion and intermodulation (IM) distortion, which have the effect of dissipating power, causing out-of-band interference, and possibly causing in-band interference with a corresponding increase in bit error rate (BER) at a receiver.
In a traditional type OFDM transmitter, a signal generator performs error correction encoding, interleaving, and symbol mapping on an input information bit sequence to produce transmission symbols. The transmission symbols are subjected to serial-to-parallel conversion at the serial-to-parallel (S/P) converter and converted into multiple parallel signal sequences. The S/P converted signal is subjected to inverse fast Fourier transform at the IFFT unit. The signal is further subjected to parallel-to-serial conversion at the parallel-to-serial (P/S) converter, and converted into a signal sequence. Then, guard intervals are added by the guard interval (GI) adding unit. The formatted signal is then up-converted to a radio frequency, amplified at the power amplifier, and finally transmitted as an OFDM signal by a radio antenna.
On the other hand, in a traditional type OFDM receiver, the radio frequency signal is down-converted to baseband or an intermediate frequency, and the guard interval is removed from the received signal at the guard interval removing unit. Then, the received signal is subjected to serial-to-parallel conversion at S/P converter, fast Fourier transform at the fast Fourier transform (FFT) unit, and parallel-to-serial conversion at P/S converter. Then, the decoded bit sequence is output.
It is conventional for each OFDM channel to have its own transmit chain, ending in a power amplifier (PA) and an antenna element. However, in some cases, one may wish to transmit two or more separate OFDM channels using the same PA and antenna, as shown in FIG. 3. This is sometimes called “carrier aggregation”. This may permit a system with additional communications bandwidth on a limited number of base-station towers. Given the drive for both additional users and additional data rate, this is highly desirable. The two channels may be combined at an intermediate frequency using a two-stage up-conversion process as shown in FIG. 3. Although amplification of real baseband signals is shown in FIG. 3, in general one has complex two-phase signals with in-phase and quadrature up-conversion (not shown). FIG. 3 also does not show the boundary between digital and analog signals. The baseband signals are normally digital, while the RF transmit signal is normally analog, with digital-to-analog conversion somewhere between these stages.
Consider two similar channels, each with average power P0 and maximum instantaneous power P1. This corresponds to a peak-to-average power ratio PAPR=P1/P0, usually expressed in dB as PAPR[dB]=10 log(P1/P0). For the combined signal, the average power is 2 P0 (an increase of 3 dB), but the maximum instantaneous power can be as high as 4 P0, an increase of 6 dB. Thus, PAPR for the combined signal can increase by as much as 3 dB. This maximum power will occur if the signals from the two channels happen to have peaks which are in phase. This may be a rare transient occurrence, but in general the linear dynamic range of all transmit components must be designed for this possibility. Nonlinearities will create intermodulation products, which will degrade the signal and cause it to spread into undesirable regions of the spectrum. This, in turn, may require filtering, and in any case will likely reduce the power efficiency of the system.
Components with required increases in linear dynamic range to handle this higher PAPR include digital-to-analog converters, for example, which must have a larger number of effective bits to handle a larger dynamic range. But even more important is the power amplifier (PA), since the PA is generally the largest and most power-intensive component in the transmitter. While it is sometimes possible to maintain components with extra dynamic range that is used only a small fraction of the time, this is wasteful and inefficient, and to be avoided where possible. An amplifier with a larger dynamic range typically costs more than one with a lower dynamic range, and often has a higher quiescent current drain and lower efficiency for comparable inputs and outputs.
This problem of the peak-to-average power ratio (PAPR) is a well-known general problem in OFDM and related waveforms, since they are constructed of multiple closely-spaced subchannels. There are a number of classic strategies to reducing the PAPR, which are addressed in such review articles as “Directions and Recent Advances in PAPR Reduction Methods”, Hanna Bogucka, Proc. 2006 IEEE International Symposium on Signal Processing and Information Technology, pp. 821-827, incorporated herein by reference. These PAPR reduction strategies include amplitude clipping and filtering, coding, tone reservation, tone injection, active constellation extension, and multiple signal representation techniques such as partial transmit sequence (PTS), selective mapping (SLM), and interleaving. These techniques can achieve significant PAPR reduction, but at the expense of transmit signal power increase, bit error rate (BER) increase, data rate loss, increase in computational complexity, and so on. Further, many of these techniques require the transmission of additional side-information (about the signal transformation) together with the signal itself, in order that the received signal be properly decoded. Such side-information reduces the generality of the technique, particularly for a technology where one would like simple mobile receivers to receive signals from a variety of base-station transmitters. To the extent compatible, the techniques disclosed in Bogucka, and otherwise known in the art, can be used in conjunction with the techniques discussed herein-below.
Various efforts to solve the PAPR (Peak to Average Power Ratio) issue in an OFDM transmission scheme, include a frequency domain interleaving method, a clipping filtering method (See, for example, X. Li and L. J. Cimini, “Effects of Clipping and Filtering on the Performance of OFDM”, IEEE Commun. Lett., Vol. 2, No. 5, pp. 131-133, May, 1998), a partial transmit sequence (PTS) method (See, for example, L. J Cimini and N. R. Sollenberger, “Peak-to-Average Power Ratio Reduction of an OFDM Signal Using Partial Transmit Sequences”, IEEE Commun. Lett., Vol. 4, No. 3, pp. 86-88, March, 2000), and a cyclic shift sequence (CSS) method (See, for example, G. Hill and M. Faulkner, “Cyclic Shifting and Time Inversion of Partial Transmit Sequences to Reduce the Peak-to-Average Ratio in OFDM”, PIMRC 2000, Vol. 2, pp. 1256-1259, September 2000). In addition, to improve the receiving characteristic in OFDM transmission when a non-linear transmission amplifier is used, a PTS method using a minimum clipping power loss scheme (MCPLS) is proposed to minimize the power loss clipped by a transmission amplifier (See, for example, Xia Lei, Youxi Tang, Shaoqian Li, “A Minimum Clipping Power Loss Scheme for Mitigating the Clipping Noise in OFDM”, GLOBECOM 2003, IEEE, Vol. 1, pp. 6-9, Dec. 2003). The MCPLS is also applicable to a cyclic shifting sequence (CSS) method.
In a partial transmit sequence (PTS) scheme, an appropriate set of phase rotation values determined for the respective subcarriers in advance is selected from multiple sets, and the selected set of phase rotations is used to rotate the phase of each of the subcarriers before signal modulation in order to reduce the peak to average power ratio (See, for example, S. H. Muller and J. B. Huber, “A Novel Peak Power Reduction Scheme for OFDM”, Proc. of PIMRC '97, pp. 1090-1094, 1997; and G. R. Hill, Faulkner, and J. Singh, “Deducing the Peak-to-Average Power Ratio in OFDM by Cyclically Shifting Partial Transmit Sequences”, Electronics Letters, Vol. 36, No. 6, 16th March, 2000).
When multiple radio signals with different carrier frequencies are combined for transmission, this combined signal typically has an increased PAPR, owing to the possibility of in-phase combining of peaks, requiring a larger power amplifier (PA) operating at low average efficiency. As taught by U.S. Pat. No. 8,582,687 (J. D. Terry), expressly incorporated herein by reference in its entirety, the PAPR for digital combinations of OFDM channels may be reduced by a Shift-and-Add Algorithm (SAA): Storing the time-domain OFDM signals for a given symbol period in a memory buffer, carrying out cyclic time shifts to transform at least one OFDM signal, and adding the multiple OFDM signals to obtain at least two alternative combinations. In this way, one can select the time-shift corresponding to reduced PAPR of the combined multi-channel signal. This may be applied to signals either at baseband, or on upconverted signals. Several decibels reduction in PAPR can be obtained without degrading system performance. No side information needs to be transmitted to the receiver, provided that the shifted signal can be demodulated by the receiver without error. This is shown schematically in FIG. 4.
Some OFDM protocols may require a pilot symbol every symbol period, where the pilot symbol may be tracked at the receiver to recover phase information (see FIG. 5). If the time-shift is performed on a given OFDM carrier, according to such a protocol, during a specific symbol period, the pilot symbol will be subject to the same time-shift, so that the receiver will automatically track these time-shifts from one symbol period to the next. However, as indicated in FIG. 2, typical modern OFDM protocols incorporate a sparser distribution of pilot symbols, with interpolation at the receiver to generate virtual pilot symbols (reference signals) for other locations. With such a protocol, an arbitrary time shift as implemented in the SAA may not be properly tracked, so that without side information, bit errors may be generated at the receiver.
What is needed is a practical method and associated apparatus for reducing the PAPR of combined OFDM signals in a wide variety of modern OFDM protocols, in a way that does not degrade the received signal or require the transmission of side-information.
The following patents, each of which are expressly incorporated herein by reference, relate to peak power ratio considerations: U.S. Pat. Nos. 7,535,950; 7,499,496; 7,496,028; 7,467,338; 7,463,698; 7,443,904; 7,376,202; 7,376,074; 7,349,817; 7,345,990; 7,342,978; 7,340,006; 7,321,629; 7,315,580; 7,292,639; 7,002,904; 6,925,128; 7,535,950; 7,499,496; 7,496,028; 7,467,338; 7,443,904; 7,376,074; 7,349,817; 7,345,990; 7,342,978; 7,340,006; 7,339,884; 7,321,629; 7,315,580; 7,301,891; 7,292,639; 7,002,904; 6,925,128; 5,302,914; 20100142475; 20100124294; 20100002800; 20090303868; 20090238064; 20090147870; 20090135949; 20090110034; 20090110033; 20090097579; 20090086848; 20090080500; 20090074093; 20090067318; 20090060073; 20090060070; 20090052577; 20090052561; 20090046702; 20090034407; 20090016464; 20090011722; 20090003308; 20080310383; 20080298490; 20080285673; 20080285432; 20080267312; 20080232235; 20080112496; 20080049602; 20080008084; 20070291860; 20070223365; 20070217329; 20070189334; 20070140367; 20070121483; 20070098094; 20070092017; 20070089015; 20070076588; 20070019537; 20060268672; 20060247898; 20060245346; 20060215732; 20060126748; 20060120269; 20060120268; 20060115010; 20060098747; 20060078066; 20050270968; 20050265468; 20050238110; 20050100108; 20050089116; and 20050089109.
The following patents, each of which is expressly incorporated herein by reference, relate to one or more topics in wireless radio-frequency communication systems: U.S. Pat. Nos. 8,130,867; 8,111,787; 8,204,141; 7,646,700; 8,520,494; 20110135016; 20100008432; 20120039252; 20130156125; 20130121432; 20120328045; 2013028294; 2012275393; 20110280169; 2013001474; 20120093088; 2012224659; 20110261676; WO2009089753; WO2013015606, 20100098139; 20130114761; WO2010077118A2.
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